Method for synchronising a signal frame of a signal transmitted from a transmitter to a receiver of a telecommunication system

ABSTRACT

The present invention relates to a method for synchronising a signal frame transmitted by a transmitter of a telecommunication system to a receiver adapted to synchronise said signal frame from a synchronisation sequence included in said signal frame. The method is characterised in that it includes: 
         a synchronisation sequence generation step ( 100 ) intended to be executed by the transmitter in the course of which the synchronisation sequence (x i (k)) is formed by the concatenation of a first and a second bursts of L complementary sequences, said first burst of L sequences being obtained by the concatenation of L times a first N pulses complementary sequence (A) of a pair of complementary sequences ((A,B)), said second burst of L sequences being obtained by the concatenation of L times a second N pulses complementary sequence (B) of said pair of complementary sequences, and    a sequence weighting step ( 200 ) intended to be executed by the transmitter in the course of which the pulses of each complementary sequence (A) of rank q of the first burst of L sequences are multiplied by the component of rank q of a L components long weighting code (MC i ) belonging to a set of I weighting codes known by the receiver (RCD) beforehand, and the pulses of each sequence of rank q of the second burst of L sequences are multiplied by the same component of rank q of said weighting code (MC i ).

The present invention relates to a method for synchronising at areceiver end a signal frame transmitted by a transmitter in atelecommunication system by transmission of a synchronisation sequenceof pulses included into said signal frame. It relates also to saidsynchronisation sequence and according to its hardware-oriented aspectsto said system, and both to a transmitter and a receiver of said system.

The transmitter is either a base station, a mobile telecommunicationdevice or any other telecommunication apparatus adapted to form and totransmit said signal frame to a receiver of the telecommunicationsystem.

The receiver is either a base station, a mobile telecommunication deviceor any other telecommunication apparatus adapted to synchronise a signalframe when it is received.

Synchronising a signal frame stands for synchronising in time andsometimes in frequency such a signal frame.

Synchronising in frequency a signal frame means that the possible phaserotation of the received signal due to a frequency drift is correctedwhen it has a significant impact on the system performance, i.e. whenlow-cost frequency oscillators are included in the receiver and/or thetransmitter. This refers in the following to the coarse frequencysynchronisation of the signal frame. The coarse frequencysynchronisation of the signal frame may not be sufficient and a residualphase rotation may remain after the received signal has been correctedby the coarse frequency synchronisation. The coarse frequencysynchronisation of the signal frame may also be non-adapted to correct alow phase rotation. Thus, synchronising in frequency a signal framemeans also that such a residual (or low) phase rotation is corrected.This refers in the following to the fine frequency synchronisation ofthe signal frame.

Synchronising in time a signal frame is usually reached by including inthe signal a synchronisation sequence which carries information fromwhich the receiver is able to get a timing reference. The receiver,which is not aware if the received signal carries such a synchronisationsequence, and does not know when the synchronisation sequence has beenincluded in the signal frame, is scanning pulse per pulse the receivedsignal continuously and gets the timing reference of a received signalframe when the received signal includes such a synchronisation sequence.

The present invention aims at solving such a synchronisation problem.

The synchronization in time of a signal frame is based, according to thepresent invention, on the perfect aperiodic autocorrelation property ofa pair of complementary sequences (A,B).

A complementary sequence A is a N bits long sequence {a₀,.a₁, . . .,a_(N−1)} such that a_(i)∈{+1,−1} and a complementary sequence B is alsoa N bits long sequence {b₀,.b₁, . . . ,b_(N−1)} such that b_(i)∈{+1,−1}.The perfect aperiodic auto-correlation property of the pair ofcomplementary sequences (A,B) is then defined by $\begin{matrix}\left\{ \begin{matrix}{{\rho_{{A,A}\quad}(k)} = {- {\rho_{B,B}(k)}}} & {\forall{k \neq 0}} \\{{\rho_{A,A}(0)} = {\rho_{B,B}(0)}} & \quad\end{matrix} \right. & (1)\end{matrix}$

where p_(A,A)(k)∀k and p_(B,B)(k)∀k are the aperiodic auto-correlationof the sequence A and B respectively defined by $\left\{ \begin{matrix}{{\rho_{A,A}(k)} = {{\sum\limits_{i = 0}^{N - k - 1}a_{i}^{*}} + a_{i + k}}} & {0 \leq k \leq {N - 1}} \\{{\rho_{B,B}(k)} = {\sum\limits_{i = 0}^{N - k - 1}{b_{i}^{*}b_{i + k}}}} & \quad\end{matrix} \right.$

By summing out the aperiodic auto-correlations p_(A,A)(k) ∀k andp_(B,B)(k) ∀k of sequences A and B, a single peak appears which isrelated to the sum of aperiodic auto-correlations p_(A,A)(k=0) andp_(B,B)(k=0). The sums of other aperiodic auto-correlations p_(A,A)(k)∀k≠0 and p_(B,B)(k) ∀k≠0 are all null. Detecting said single peak leadsto a precise synchronisation in time of a signal frame, i.e. thereceiver is then able to define the beginning of the signal frame whichis being received.

Note that equation (1) provides the definition of a pair of bi-polarcomplementary sequences which is used in the following, but because sucha definition may be generalised easily to non-binary sequences, thescope of the present invention is not limited to bi-polar complementarysequences.

Golay sequences, defined for example in the book “Golay ComplementarySequences” (M. G. Parker et al., june 2004 M. G. Parker, K. G. Patersonand C. Tellambura, Wiley Encyclopedia of Telecommunications, Editor: J.G. Proakis, Wiley Interscience, 2002), are an example of such a pair ofcomplementary sequences.

Indeed, the present invention aims at providing a method forsynchronising a signal frame transmitted by a transmitter of atelecommunication system to a receiver adapted to synchronise saidsignal frame from a synchronisation sequence included in said signalframe, characterised in that it includes a synchronisation sequencegeneration step intended to be executed by the transmitter in the courseof which the synchronisation sequence is formed by the concatenation ofa first and a second bursts of L complementary sequences, said firstburst of L sequences being obtained by the concatenation of L times afirst N pulses complementary sequence of a pair of complementarysequences, said second burst of L sequences being obtained by theconcatenation of L times a second N pulses complementary sequence ofsaid pair of complementary sequences.

The method for synchronising further includes a sequence weighting stepintended to be executed by the transmitter in the course of which thepulses of each complementary sequence of rank q of the first burst of Lsequences are multiplied by the component of rank q of a L componentslong weighting code belonging to a set of I weighting codes known by thereceiver beforehand, and the pulses of each sequence of rank q of thesecond burst of L sequences are multiplied by the same component of rankq of said weighting code.

An advantage of the present invention is that each L long componentsweighting code may be representative of any information which shall betransmitted from the transmitter to the receiver. For example, when sucha synchronisation method is included in a cellular telecommunicationsystem, the weighting code may be representative of a cell or cellsector characteristic, such as an identifier, or an information whichallows to get such a cell or cell sector characteristic.

According to an aspect of the present invention, in the course of thesynchronisation sequence generation step at least one guard interval,defined by a W bits long cyclic extension of either the first or secondcomplementary sequence is included in the synchronisation sequence. Saidat least one guard interval is located either at the beginning or at theend of the first burst of L sequences when said at least one guardinterval is defined from the first complementary sequence, and said atleast one guard interval is located either at the beginning or at theend of the second burst of L sequences when said at least one guardinterval is defined from the second complementary sequence.

According to an embodiment of the synchronisation sequence generationstep, two guard intervals are defined by W bits long cyclic extensionsof the first complementary sequence. One of said guard interval islocated at the beginning of the first burst of L sequences and the otherone is located at the end of the first burst of L sequences. Moreover,two guard intervals are defined by W bits long cyclic extensions of thesecond complementary sequence. One of said guard interval is located atthe beginning of the second burst of L sequences and the other one islocated at the end of the second burst of L sequences.

In the following the receiver is considered as being equipped by atleast one antenna intended each to receive said signal frame.

According to an embodiment of the present invention, the set of Iweighting codes is a set of orthogonal codes from which a fast transformis obtained. Each weighting code is a Hadamard code according to a firstexample and a Hadamard fast transform is obtained. According to an otherexample each weighting code is a Fourier code and a fast Fouriertransform is obtained.

Using orthogonal codes ensures orthogonality between information carriedby such modulation codes. This property is particularly interesting inthe case of a cellular telecommunication system when the weighting codesare representative of cell or sector identifiers because it ensuresorthogonality between neighboring cells or cell sectors.

According to a variant of this embodiment, the weighting codes aremultiplied by a same scrambling code which is, for example, a Barkercode.

According to another aspect of the present invention, the method forsynchronising is characterised in that it further includes for each ofsaid at least one antenna, a resulting stream computation step in thecourse of which at least one resulting stream is obtained from thesignal which is being received by the antenna, said resulting streamcomputation step includes

-   -   at least one time-shifting sub-step in the course of each a        time-delayed version of the received signal is computed,    -   at least one first sequence correlation sub-step in the course        of each a first correlation stream is computed by correlating        the time-delayed version of the received signal with a replica        of the first complementary sequence of the pair of complementary        sequences,    -   at least one second sequence correlation sub-step in the course        of each a second correlation stream is computed by correlating        the received signal with a replica of the second complementary        sequence of the pair of complementary sequences, and    -   at least one stream summation sub-step in the course of each a        resulting stream is formed from the first correlation stream and        the second correlation stream.

Moreover, the method for synchronising includes also an output streamcomputation step in the course of which at least one output stream isobtained for each of said at least one resulting stream and a timeinstant determination and weighting code retrieval step in the course ofwhich at least one decision value at a time instant is computed fromsaid at least one output stream. A time instant, at which the signalframe which is being received is said synchronised in time, is thenobtained by maximising said at least one decision value, and theweighting code carried by the received synchronisation sequence isretrieved from L peaks separated by N positions from each other of atleast one of said resulting streams obtained from said maximum decisionvalue. The first of said L peaks of each of said at least one resultingstream is located at the time instant at which the signal frame is saidsynchronised in time.

A resulting stream which carries a single peak from which a signal frameis synchronised in time exhibits possibly secondary peaks which are dueto bad autocorrelation properties of weighting codes. This drasticallyincreases the detection error probability. In order to counter fightthis problem, it is advantageous that a scrambling code is used in orderto bound the norms of said secondary peaks.

According to a first embodiment of the resulting stream computation steprelated to an antenna, the resulting stream computation step includes

-   -   a single time-shifting sub-step in the course of which a        time-delayed version of the received signal is computed,    -   a single first sequence correlation sub-step in the course of        which a first correlation stream is computed by correlating the        time-delayed version of the received signal with a replica of        the first complementary sequence of the pair of complementary        sequences,    -   a single second sequence correlation sub-step in the course of        which a second correlation stream is computed by correlating the        received signal with a replica of the second complementary        sequence of the pair of complementary sequences, and    -   a single stream summation sub-step in the course of which a        resulting stream is formed by summing said first correlation        stream and said second correlation stream.

According to a second embodiment of the resulting stream computationstep related to an antenna, M frequency offset values being predefinedby the receiver from a range of possible frequency drifts, the resultingstream computation step includes

-   -   a single time-shifting sub-step in the course of which a        time-delayed version of the received signal is computed,    -   a single first sequence correlation sub-step in the course of        which a first correlation stream is computed by correlating the        time-delayed version of the received signal with a replica of        the first complementary sequence of the pair of complementary        sequences,    -   a single second sequence correlation sub-step in the course of        which a second correlation stream is computed by correlating the        received signal with a replica of the second complementary        sequence of the pair of complementary sequences,    -   M coarse sequence by sequence frequency correction sub-steps in        the course of each the phase of pulses of said second        correlation stream related to a same complementary sequence is        corrected by a constant value related to one of said M frequency        offset values, and    -   M stream summation sub-steps in the course of each a resulting        stream is formed by summing the first correlation stream and one        of said M corrected second correlation streams.

This embodiment is advantageous because the application of the coarsefrequency correction on the second correlation stream rather than on thefull received signal provides a lower complexity receiver.

According to a third embodiment of the resulting stream computation steprelated to an antenna, M frequency offset values being predefined by thereceiver from a range of possible frequency drifts, the resulting streamcomputation step includes

-   -   M coarse pulse by pulse frequency correction sub-steps in the        course of each the phase of each pulse of the received signal is        corrected by a linearly increasing value the slope of which is        related to one of said M frequency offset values,    -   M time-shifting sub-steps in the course of each a time-delayed        version of one of said M received and phase corrected signals is        computed,    -   M first sequence correlation sub-steps in the course of each a        first correlation stream is computed by correlating the        time-delayed version of one of said M received and phase        corrected signals with a replica of the first complementary        sequence A of the pair of complementary sequences,    -   M second sequence correlation sub-steps in the course of each a        second correlation stream is computed by correlating one of said        M received and phase corrected signals with a replica of the        second complementary sequence of the pair of complementary        sequences, and    -   M stream summation sub-steps in the course of each a resulting        stream is formed from one of said M first correlation streams        and one of said M second correlation streams related to the same        received and phase corrected signal.

According to a first embodiment of the time instant determination andweighting code retrieval step, at a time instant, a decision value iscomputed for each of said at least one output stream obtained from eachof said at least one antenna.

According to a second embodiment of the time instant determination andweighting code retrieval step, at a time instant, a decision value iscomputed for at least one combination of said at least one outputstream. Each of said at least one combination, which is related toeither one of said I weighting codes or one of said M frequency offsetsand one of said I weighting codes, is defined by the square root of thesum of squared modules of output streams obtained from said at least oneantenna which are related to either a same weighting code or the sameweighting code and the same frequency value.

According to a first embodiment of the computation of a decision value,each decision value computed at a time instant related to an outputstream, respectively a combination of output streams, is the squarednorm of said output stream, respectively the combined output stream,evaluated at said time instant.

According to second embodiment of the computation of a decision value,each decision value computed at a time instant related to an outputstream, respectively a combination of output streams, is a correlationmerit factor defined by the ratio of the energy at said time instant ofsaid output stream, respectively said combination of output streams,divided by the energy of said output stream, respectively saidcombination of output streams, averaged on two time intervals definedrespectively before and after said time instant.

According to a first embodiment of the output stream computation step, Ioutput streams are obtained for each of said at least one resultingstream, each of said I output streams related to one of said at leastone resulting stream is obtained by correlating, at a given timeinstant, said resulting stream with a comb of pulses related to one ofsaid I weighting codes. Each pulse of a comb related to a weighting codeis separated to each other by N positions. Moreover, each pulse of saidcomb is weighted by a component of said weighting code.

According to a second embodiment of the output stream computation step,the I weighting codes being orthogonal to each other, a fast transformrelated to said orthogonal weighting codes being obtained, I outputstreams are obtained for each of said at least one resulting stream,each of said I output streams related to each of said at least oneresulting stream is obtained by processing L pulses of said resultingstream separated to each other by N positions with said fast transform.

According to a third embodiment of the output stream computation step, asingle output stream is obtained for each of said at least one resultingstream by computing, at a time instant, the squared root of the sum ofthe energy of L pulses separated by N positions from each other of saidresulting stream.

This embodiment is advantageous because it provides a lower complexityreceiver compared to the complexity of the receiver defined according toprevious embodiments. Such a low complexity receiver is due to the factthat the location of the synchronisation sequence power is looking forrather than performing multiple correlations as above-described inprevious embodiments.

When the output streams are obtained from the third embodiment of theoutput stream computation step, each of (L−1) components following thefirst component of each of said I weighting codes being differentiallyencoded from its first component, the signal frame being possiblycorrected in frequency, L peaks separated by N positions from each otherof a single resulting stream obtained from said maximal decision valueand related to each of said at least one antenna being considered, thefirst of said L peaks being located at the time instant at which thesignal frame is synchronised, in the course of the time instantdetermination and weighting code retrieval a soft estimate of acomponent of rank q of the weighting code carried by the received signalframe is obtained from the product of the sum of the phase rotation andamplitude corrected peak of rank q of said at least one single resultingstream by a weighting value.

According to a first embodiment of the phase rotation and amplitude peakcorrection, the phase rotation and amplitude of said at least one peakof rank q are corrected by multiplying said peak of rank q by thecomplex conjugate of the peak preceding said (L−1) peaks, and saidweighting value is the square root of the sum of the square of themodule of peaks preceding the (L−1) peaks of each of said at least onesingle resulting stream.

According to a second embodiment of the phase rotation and amplitudepeak correction, the phase rotation and amplitude of said at least onepeak of rank q are corrected by multiplying said at least one peak ofrank q by the product of the complex conjugate of the peak precedingsaid (L−1) peaks divided by its module by the square root of the averageenergy of said L peaks, and said weighting value is the square root ofthe sum of the average energies of said L peaks obtained for each ofsaid at least one single resulting stream.

According to another aspect of the present invention, the signal framereceived by each of said at least one antenna being transmitted on amultipath channel having P consecutive paths and being synchronised intime and possibly coarsely corrected in frequency, in the course of thetime instant determination and weighting code retrieval step the phaseof each pulse of a received signal frame is corrected by a linearlyincreasing value the slope of which is obtained from a weighted averageof at least P slope estimations obtained for each of said at least oneantenna. Each of said P slope estimations is weighted by an estimate ofthe squared amplitude of one of said P path coefficients, and theweighting value of said average is the sum of the squared amplitude ofsaid P path coefficients.

According to an embodiment for computing a slope estimation, each slopeestimation related to a path coefficient is obtained by the ratio of asequence by sequence slope estimation related to said path coefficientover N, said sequence by sequence slope estimation being defined by theaverage of (L−1) differences between the phase of the sum of a first anda second correlation factor related to said path coefficient andcomputed at a first time instant on a segment of respectively a firstand second complementary sequence of the received synchronisationsequence and the phase of the sum of a first and a second correlationfactor related to said path coefficient and computed at a second timeinstant on a segment of respectively a first and second complementarysequence of the received synchronisation sequence.

According to another aspect of the present invention, the signal istransmitted from multiple antennas. The method for synchronisingaccording to the present invention is then characterised in that thetransmitter transmits the same signal from each of said antennas withminor different time delays.

According to one of its hardware oriented aspects, the present inventionrelates to a transmitter intended to execute the above-mentionedsynchronisation sequence generation and sequence weighting steps.

According to another of its oriented aspects, the present inventionrelates to a receiver intended to execute the above-mentioned resultingstream computation step for each of its antennas, and theabove-mentioned output stream computation step and the above-mentionedtime instant determination and weighting code retrieval step.

The characteristics of the invention mentioned above, as well as others,will emerge more clearly from a reading of the following descriptiongiven in relation to the accompanying figures, amongst which:

FIG. 1 represents a synoptic schema of an example of telecommunicationsystem according to the present invention,

FIG. 2 is a diagram which represents the steps of a method forsynchronising a signal frame of a received signal intended to beexecuted by the transmitter according to the present invention,

FIG. 3 represents an example of a synchronisation sequence according tothe present invention,

FIG. 4 depicts the transmission of a synchronisation sequence on amultipath transmission channel,

FIG. 5 represents a diagram which represents the steps of a method forsynchronising a signal frame of a received signal intended to beexecuted by the receiver according to the present invention,

FIG. 6 represents a diagram of a first embodiment of the resultingstream computation step,

FIG. 7 represents a diagram of a second embodiment of the resultingstream computation step,

FIG. 8 represents a diagram of a third embodiment of the resultingstream computation step,

FIG. 9 represents a diagram of a first embodiment of the output streamcomputation step,

FIG. 10 represents a diagram of a second embodiment of the output streamcomputation step,

FIG. 11 represents a diagram of a third embodiment of the output streamcomputation step,

FIG. 12 represents a chronogram which depicts the computation of adecision value,

FIG. 13 depicts how correlation factors are computed,

FIG. 14 represents a schema of a transmitter according to the presentinvention,

FIG. 15 represents a schema of a receiver according to the presentinvention, and

FIG. 16 a to 16 c depicts different embodiments of the resulting streamcombiner includes in the receiver.

FIG. 1 represents a synoptic schema of an example of a telecommunicationsystem SYST according to the present invention. A radiotelecommunication is depicted in FIG. 1 but the present invention is notrestricted to such telecommunication systems. The telecommunicationsystem SYST includes a receiver RCD which is intended to receivecontinuously a signal from N_(ant) antennas RANT_(s) and othercommunications means not depicted. When such a received signal is asignal Tsg(k) which is generated and transmitted by the transmitter TRDof the telecommunication system SYST, from the antenna TANT (and othercommunication means not depicted) to the receiver RCD, each signal frameof the signal Tsg(k) is being synchronised at the receiver end.

FIG. 2 is a diagram which represents the steps of a method forsynchronising a signal frame of the received signal Tsg(k) intended tobe executed by the transmitter TRD according to the present invention.

The synchronisation method includes a synchronisation sequencegeneration step 100 and a sequence weighting step 200. In the course ofthe synchronisation sequence generation step 100 a synchronisationsequence x^(i)(k) is embedded in each signal frame of the signal Tsg(k).The synchronisation sequence x^(i)(k) is generated from a pair ofcomplementary sequences (A,B) defined in the opening paragraph. In thecourse of the sequence weighting step 200, the generated synchronisationsequence x^(i)(k) is weighted by a weighting code MC¹={c₀ ^(i), . . .,c₁ ^(i), . . . ,c_(L−1) ^(i)} which belongs to a set of I weightingcodes MC^(i),i∈{1, . . . ,I} known by the receiver beforehand.

In the course of the synchronisation sequence generation step 100, asynchronisation sequence x^(i)(k) is formed by the concatenation of afirst and a second bursts of L complementary sequences. The first burstof L sequences includes L times a first complementary sequence A of thepair of complementary sequences (A,B), and the second burst of Lsequences includes L times the second complementary sequence B of thepair of complementary sequences (A,B). The first and second bursts of Lsequences include then each L.N pulses.

In the course of the sequence weighting step 200, the pulses of eachcomplementary sequence A of rank q of the first burst of L sequences aremultiplied by the component of rank q of a L components weighting codeMC^(i)={c₀ ^(i), . . . ,c₁ ^(i), . . . ,c_(L−1) ^(i)}, and the pulses ofeach complementary sequence B of rank q of the second burst of Lsequences are multiplied by the same component of rank q of saidweighting code.

According to an aspect of the present invention, in the course of thesynchronisation sequence generation step 100, at least one guardinterval, defined by a W bits long cyclic extension of either the firstor second complementary sequence is included in the synchronisationsequence x^(i)(k). Said at least one guard interval is located either atthe beginning or at the end of the first burst of L sequences when saidat least one guard interval is defined from the first complementarysequence A, and said at least one guard interval is located either atthe beginning or at the end of the second burst of L sequences when saidat least one guard interval is defined from the second complementarysequence B.

According to an embodiment of the synchronisation sequence generationstep 100, four guard periods Δ_(A) ^(Le), Δ_(A) ^(Ri), Δ_(B) ^(Le) andΔ_(B) ^(Ri) are defined. The index ‘Le’ and ‘Ri’ means that the cyclicextension is located respectively at the beginning and the end of aburst of L sequences. More precisely, the cyclic extension Δ_(A) ^(Le),which is obtained from a W long cyclic extension of the complementarysequence A, is included at the beginning of the first burst of Lsequences, the cyclic extension Δ_(A) ^(Ri), which is also obtained froma W long cyclic extension of the complementary sequence A, is includedat the end of the first burst of L sequences, the cyclic extension Δ_(B)^(Le), which is obtained from a W long cyclic extension of thecomplementary sequence B, is included at the beginning of the secondburst of L sequences, and the cyclic extension Δ_(B) ^(Ri), which isalso obtained from a W long cyclic extension of the complementarysequence B, is included at the end of the second burst of L sequences.FIG. 3 shows an example of such a synchronisation sequence x^(i)(k).

The synchronisation sequence x^(i)(k) at the transmitter output is thusgiven by the general equation (2) in which the cyclic extension value Wmay be null if the bursts of L sequences are not combined with guardintervals: $\begin{matrix}{{x^{i}(k)} = {{\Delta_{A}^{Le}(k)} + {\sum\limits_{l = 0}^{L - 1}{c_{l}^{i} \cdot {A\left\lbrack {k - W - {l \cdot N}} \right\rbrack}}} + {\Delta_{A}^{Ri}\left( {k - W - {N \cdot L}} \right)} + {\Delta_{B}^{Le}\left( {k - u} \right)} + {\sum\limits_{l = 0}^{L - 1}{c_{l}^{i} \cdot {B\left\lbrack {k - W - {l \cdot N} - u} \right\rbrack}}} + {\Delta_{B}^{Ri}\left( {k - W - {N \cdot L} - u} \right)}}} & (2)\end{matrix}$

with u=N.L+2.W and A[k]=B[k]=0 for k<0 and k≧N.

In the telecommunication system SYST, the signal Tsg(k) flows betweenthe transmitter TRD and the receiver RCD on a multipath transmissionchannel CH. In the case of a radio telecommunication system for example,the transmission channel CH is called multipath because between thetransmitter TRD and the receiver RCD there are usually obstacles onwhich waves are reflected. Then, a transmitted signal Tsg(k) ispropagating along several paths between the transmitter TRD and thereceiver RCD.

FIG. 4 depicts the transmission of the synchronisation sequence x^(i)(k)to an antenna RANT_(s) on a multipath transmission channel CH having Pconsecutive paths. The model of such a channel CH is then a P branchesmodel. The synchronisation sequence X^(i)(k) is thus transmitted on eachof P branches by first delaying it by p pulses and multiplying it by acomplex path coefficient h_(p,s). Next, due to some imperfection of thetelecommunications means of the receiver RCD, a random signal n(k) whichis usually modeled by a white Gaussian noise, is added to the sum of Pterms X^(i)(k−p).h_(p,s)(k).

Consequently, the synchronisation sequence y_(s)(k) which is possiblycarried by a signal received by one of antennas RANT_(s) of the receiverRCD is given by: $\begin{matrix}{{y_{s}(k)} = {\left\lbrack {\sum\limits_{p = 0}^{P - 1}{{x^{i}\left( {k - p} \right)} \cdot {h_{p,s}(k)}}} \right\rbrack + {n(k)}}} & (3)\end{matrix}$

where the index s refers to the antenna RANT_(s).

When the signal which is being received is the signal Tsg(k), thesynchronisation sequence y_(s)(k) is given by equation (4) once equation(2) and (3) are combined: $\begin{matrix}{{y_{s}(k)} = {{\sum\limits_{p = 0}^{P - 1}{\begin{bmatrix}{{\Delta_{A}^{Le}\left( {k - p} \right)} + {\sum\limits_{l = 0}^{L - 1}{c_{l}^{i} \cdot {A\left\lbrack {k - p - W - {l \cdot N}} \right\rbrack}}} +} \\{{\Delta_{A}^{Ri}\left( {k - p - W - {N \cdot L}} \right)} +} \\{{\Delta_{B}^{Le}\left( {k - p - u} \right)} + {\sum\limits_{l = 0}^{L - 1}{c_{l}^{i} \cdot {B\left\lbrack {k - p - W - {l \cdot N} - u} \right\rbrack}}} +} \\{\Delta_{B}^{Ri}\left( {k - p - W - {N \cdot L} - u} \right)}\end{bmatrix} \cdot {h_{p,s}(k)}}} + {n(k)}}} & (4)\end{matrix}$

FIG. 5 shows a diagram which represents the steps of the method forsynchronising a signal frame of a received signal intended to beexecuted by the receiver RCD according to the present invention. Themethod for synchronising includes, a resulting stream computation step300 _(s) for each antenna RANT_(s) of the receiver RCD, an output streamcomputation step 400 and a time instant determination and weighting coderetrieval step 500.

In the course of each resulting stream computation step 300 _(s) atleast one resulting streams is obtained from the signal which is beingreceived by the antenna RANT_(s) and which possibly carries thesynchronisation sequence y_(s)(k).

In the course of the output stream computation step 400 at least oneoutput stream is obtained for each of said at least one resulting streamand in the course of the time instant determination and weighting coderetrieval step 500 at least one decision value at a time instant k iscomputed from said at least one output stream. A time instant k_(best),at which the signal frame which is being received is said synchronisedin time, is then obtained by maximising said at least one decisionvalue. When the synchronisation sequence y_(s)(k) is being received, theweighting code MC^(j) ^(best) (j_(best)∈{1, . . . ,I}) carried by saidreceived synchronisation sequence y_(s) (k) is retrieved from L peaksseparated by N positions from each other of at least one of saidresulting streams obtained from said maximum decision value. The firstof said L peaks of each of said at least one resulting streams is thenlocated at the time instant k_(best).

FIG. 6 represents a diagram of a first embodiment of the resultingstream computation step 300 _(s) related to an antenna RANT_(s). Theresulting stream computation step 300 _(s) includes a singletime-shifting sub-step 301 _(s), a single first sequence correlationsub-step 302 _(s), a single second sequence correlation sub-step 303_(s), and a single stream summation sub-step 304 _(s).

In the course of the time-shifting sub-step 301 _(s), the receivedsignal is time-delayed by a time duration u. The time duration u isdefined according to the generated synchronisation sequence x^(i)(k)which is possibly carried by the received signal. For example, when thesynchronisation sequence x^(i)(k) is given by equation (2), the timeduration u is equal to (L.N+2.W) times the duration between toconsecutive pulses of the synchronisation sequence x^(i)(k), andaccording to another example, it is equal to (L.N) times said durationwhen the synchronisation sequence x^(i)(k) is not combined with cyclicextensions.

In the course of the first sequence correlation sub-step 302 _(s), afirst correlation stream y_(A,s)(k) is computed by correlating thetime-delayed version of the received signal with a replica of the firstcomplementary sequence A of the pair of complementary sequences (A,B).

In the course of the second sequence correlation sub-step 303 _(s), asecond correlation stream y_(B,s)(k) is computed by correlating thereceived signal with a replica of the second complementary sequence B ofthe pair of complementary sequences (A,B).

When the synchronisation sequence y_(s)(k) is carried by the receivedsignal, the first correlation stream y_(A)(k) is given by:$\begin{matrix}{{y_{A,s}(k)} = {\left\lbrack {{\sum\limits_{p = 0}^{P - 1}{{x^{i}\left( {k - p - u} \right)} \cdot {h_{p,s}(k)}}} + {n\left( {k - u} \right)}} \right\rbrack*{A^{*}\left( {- k} \right)}}} & (5)\end{matrix}$and the second correlation stream y_(B,s)(k) is given by:$\begin{matrix}{{y_{B,s}(k)} = {\left\lbrack {{\sum\limits_{p = 0}^{P - 1}{{x^{i}\left( {k - p} \right)} \cdot {h_{p,s}(k)}}} + {n(k)}} \right\rbrack*{B^{*}\left( {- k} \right)}}} & (6)\end{matrix}$

In the course of the stream summation sub-step 304 _(s), a resultingstream r_(s)(k) is formed by summing the two correlation streamsy_(A,s)(k) and y_(B,s)(k).

When the synchronisation sequence y_(s)(k) is carried by the receivedsignal, the L sequences A of the first burst of L sequences are thentime aligned with the L sequences B of the second burst of L sequences.The resulting stream r_(s)(k) is then given by $\begin{matrix}{{{r_{s}(k)} = {{{y_{A,s}(k)} + {y_{B,s}(k)}} = {{\sum\limits_{p = 0}^{P - 1}{{\beta^{i}\left( {k - p} \right)} \cdot {h_{p,s}(k)}}} + {\alpha(k)}}}}{{{where}\quad{\alpha(k)}} = {{{n\left( {k - u} \right)}*{A^{*}\left( {k - u} \right)}} + {{n(k)}*{B^{*}\left( {- k} \right)}\quad{and}}}}\begin{matrix}{{\beta^{i}(k)} = {{{x^{i}\left( {k - u} \right)}*{A^{*}\left( {- k} \right)}} + {{x^{i}(k)}*{B^{*}\left( {- k} \right)}}}} \\{= {\begin{bmatrix}{{{\Delta_{A}^{Le}\left( {k - u} \right)}*{A^{*}\left( {- k} \right)}} +} \\{{\Delta_{B}^{Le}\left( {k - u} \right)}*{B^{*}\left( {- k} \right)}}\end{bmatrix} + {\sum\limits_{l = 0}^{L - 1}{c_{l}^{i} \cdot 2 \cdot N \cdot {\delta\left( {k - W - {l \cdot N} - u} \right)}}} +}} \\{\begin{bmatrix}{{{\Delta_{A}^{Ri}\left( {k - W - {N \cdot L} - u} \right)}*{A^{*}\left( {- k} \right)}} +} \\{{\Delta_{B}^{Ri}\left( {k - W - {N \cdot L} - u} \right)}*{B^{*}\left( {- k} \right)}}\end{bmatrix}}\end{matrix}} & (7)\end{matrix}$

Thus, according to this embodiment, at the output of the resultingstream computation step 300 _(s) a single resulting stream r_(s)(k)isobtained for each antenna RANT_(s).

As mentioned in the opening paragraph, the frequency drift between thetransmitter TRD and the receiver RCD may have a significant impact onthe system performance. The phase of the received signal shall then becorrected in order to avoid that said frequency drift decreases theprobability for detecting carried data and to degrade the aperiodicauto-correlation properties of complementary sequences A and B (given byequation (1)) carried by the received signal Tsg(k). Such a degradationis due to, first, the linearly varying phase rotation from pulse topulse inside each complementary sequence itself, which destroys saidproperties of each sequence with its local replica, and, second, theintegration of this pulse to pulse phase rotation on the first half ofthe synchronisation sequence x^(i)(k) which leads to an important phaseoffset between the complementary sequences of a pair. The worst caseoccurs when the phase offset equals π since then the amplitude of thesingle peak is null and the amplitudes of aperiodic auto-correlationsp_(A,A)(k) and p_(B,B)(k) ∀k≠0 add to each other.

The receiver RCD is not aware of the precise value of the frequencydrift but it knows beforehand the range to which said frequency driftbelongs. The receiver determines then M possible frequency offset valuesΔf_(m) m={1, . . . ,M} from said range. Note that one of possiblefrequency offset values Δf_(m) may be equal to 0 if the receiver is notaware if a frequency drift occurs, i.e. in order to consider also theversion of the synchronisation sequence without any frequencycorrection.

FIG. 7 represents a diagram of a second embodiment of the resultingstream computation step 300 _(s) related to an antenna RANT_(s) whichprovides a first way for coarsely correcting in frequency the receivedsignal.

The resulting stream computation step 300 _(s) includes a singletime-shifting sub-step 301 _(s) a single first sequence correlationsub-step 302 _(s), and a single second sequence correlation sub-step 303_(s) in order to obtain the two correlation streams y_(A,s)(k) andy_(B,s)(k) as above-explained.

Moreover, the resulting stream computation step 300 _(s) includes Mcoarse sequence by sequence frequency correction sub-steps 305 _(s,m)and M stream summation sub-steps 304 _(s,m).

In the course of each sequence by sequence frequency correction sub-step305 _(s,m) the phase of pulses of the second correlation streamy_(B,s)(k) related to a same complementary sequence is corrected by aconstant value given byΔΦ_(m)=Δφ_(m)(N.L+2.W)

where Δφ_(m)=2π.Δf_(m).T_(sa) is the phase rotation between twoconsecutive pulses separated by $T_{sa} = \frac{1}{f_{sa}}$with f_(sa) the sampling frequency and Δf_(m) a possible frequencyoffset value. A corrected second correlation stream y_(B,s,m)(k) is thenobtained.

In the course of each stream summation sub-step 304 _(s,m,) a resultingstream r_(s,m)(k) is formed by summing the first correlation streamy_(A,s)(k) and the corrected second correlation stream y_(B,s,m)(k)Thus, according to this embodiment, at the output of the resultingstream computation step 300 _(s) M resulting streams r_(s,m)(k)areobtained for each antenna RANT_(s).

FIG. 8 represents a diagram of a third embodiment of the resultingstream computation step 300 _(s) related to an antenna RANT_(s) whichprovides a second way for coarsely correcting in frequency the receivedsignal.

The resulting stream computation step 300 _(s) includes M branches eachof which includes the same sub-steps from which a resulting streamr_(s,m)(k) is obtained.

A branch m includes a coarse pulse by pulse frequency correctionsub-step 306 _(s,m) in the course of which the phase of each pulse ofthe received signal is corrected by a linearly increasing value theslope of which is Δφ_(m). A corrected received signal y_(s,m)(k) is thenobtained.

The branch m includes also a single time-shifting sub-step 301 _(s,m), asingle first sequence correlation sub-step 302 _(s) and a single secondsequence correlation step 303 _(s,m) in order to obtain the twocorrelation streams y_(A,s,m)(k) and y_(B,s,m)(k) from the correctedreceived signal y_(s,m)(k). It further includes a stream summationsub-step 304 _(s,m) in the course of which a resulting stream r_(s,m)(k)is formed by summing the first correlation stream y_(A,s,m)(k) and thesecond correlation stream y_(B,s,m)(k).

Thus, according to this embodiment, at the output of the resultingstream computation step 300 _(s) M resulting streams r_(s,m)(k) areobtained for each antenna RANT_(s).

As above-explained, at the output of the resulting stream computationstep 300 _(s), either a single resulting stream r_(s)(k) is obtained orM resulting streams r_(s,m)(k) are obtained. In the following, a generalcase is considered in which M resulting streams r_(s,m)(k) are obtained.Simpler cases can be deduced from such a general case. For example, asingle resulting stream r_(s)(k) is obtained from this general case whenM=1. Moreover, when no coarse frequency correction is required, a singleresulting stream is obtained for Δf_(m)=0 and M=1.

FIG. 9 represents a diagram of a first embodiment of the output streamcomputation step 400.

According to this first embodiment I correlation output streams z_(s,m)^(j)(k) are computed for each of M resulting streams r_(s,m)(k) obtainedfrom the resulting stream computation step 300 _(s). An output streamz_(s,m) ^(j)(k) related to a resulting stream r_(s,m)(k) is obtained bycorrelating at a given time instant k the resulting stream r_(s,m)(k)with a comb of L pulses cp^(j) related to one of said I weighting codesMC^(j). Each pulse of the comb of pulses cp^(j) is separated to eachother by N positions and each pulse q of the comb of pulses cp^(j) isweighted by a component c_(q) ^(j) of a weighting code MC^(j).

FIG. 10 represents a diagram of a second embodiment of the output streamcomputation step 400.

According to this second embodiment, the set of I weighting codes MC^(j)is a set of orthogonal codes from which a fast transform TRANS isobtained. Each of said weighting codes is a Hadamard code according to afirst example, and a Fourier code according to a second example.

In the course of the output stream computation step 400, I outputstreams z_(s,m) ^(j)(k) are computed for each of M resulting streamsr_(s,m)(k) obtained from the resulting stream computation step 300 _(s).Each of I output streams z_(s,m) ^(j)(k) related to a resulting streamr_(s,m)(k) is obtained by processing L pulses r_(s,m)(k), r_(s,m)(k+N),. . . , r_(s,m)(k+(L−1).N), selected from the resulting streamr_(s,m)(k) and separated to each other by N positions, with the fasttransform TRANS obtained from the I orthogonal weighting codes MC^(j).

Thus, according to these two embodiments of the output streamcomputation step 400, I output streams z_(s,m) ^(j)(k) are obtained foreach resulting stream r_(s,m)(k).

FIG. 11 represents a diagram of a third embodiment of the output streamcomputation step 400.

According to this embodiment, a single output stream z_(s,m)(k) iscomputed for each of M resulting streams r_(s,m)(k) obtained from theresulting stream computation step 300 _(s).

The single output stream z_(s,m)(k) is given at a time instant k by${z_{s,m}(k)} = \sqrt{\sum\limits_{\ell = 0}^{L - 1}{{r_{s,m}\left( {k + {\ell \cdot N}} \right)}}^{2}}$

where |r_(s,m)(k+l.N)|² is the energy of a pulse separated by l.Npositions from the first of L pulses. Said L pulses r_(s,m)(k),r_(s,m)(k+N), . . . , r_(s,m)(k+(L−1).N)are selected from the resultingstream r_(s,m)(k) and separated to each other by N positions.

When the received signal is not the signal Tsg(k) transmitted by thetransmitter TRD, none of output streams z_(s,m) ^(j)(k), respectivelyz_(s,m)(k), obtained at the output of the output stream computation step400 if one of the two first embodiments, respectively the thirdembodiment, of the output stream computation step 400 is used, carries apeak which can be considered as the single peak from which thesynchronisation in time of a signal frame is reached.

But, when the received signal is the signal Tsg(k), such a single peakappears at the time k_(best) in at least one of output streams z_(s,m)^(j) ^(best) (k), respectively z_(s,m)(k). The signal frame which isbeing received is then said synchronised in time and, as mentioned inthe opening paragraph, the receiver is able to determine precisely thebeginning of said signal frame and thus the beginning of thesynchronisation sequence x^(j)(k) carried by said signal frame.

Note that the weighting code MC^(j) ^(best) which is carried by thereceived synchronisation sequence y_(s)(k) is deduced directly from thestream z_(s) _(best) _(,m) _(best) ^(j) ^(best) (k) obtained from theantenna RANT_(sbest) following the maximisation of the decision valuedescribed later.

Mathematically speaking, such a single peak appears in a time windowwhich is centered on the time instant k_(best) and which is of durationu equal in the general case to (L.N+2.W) because the coefficient β^(j)^(best) (k) over such a time window may be rewritten as $\begin{matrix}{{\beta_{\lbrack u\rbrack}^{j_{best}}(k)} = {\sum\limits_{l = 0}^{L - 1}\quad{c_{l}^{j_{best}}{{.2}.N.{\delta\left( {k - W - {l.N} - u} \right)}}}}} & (8)\end{matrix}$

An output stream z_(s,m) ^(j) ^(best) (k) in which the single peakappears is then given by: $\begin{matrix}{{z_{s,m}^{j_{best}}(k)} = {\sum\limits_{n = 0}^{L - 1}\quad{{r_{\lbrack u\rbrack}\left( {k + {n.N}} \right)}.c_{n}^{*_{j_{best}}}}}} & (9)\end{matrix}$

By including equation (7) in which the sequences β^(j) ^(best) (k) isgiven by equation (8) in equation (9), and following some mathematicalmanipulations, said correlation output stream z_(s,m) ^(j) ^(best) (k)is given by${z_{s,m}^{j_{best}}(k)} = {{\sum\limits_{p = 0}^{P - 1}\quad{\sum\limits_{l = 0}^{L - 1}\quad{\sum\limits_{n = 0}^{L - 1}\quad{{c_{l}^{j_{best}}.c_{n}^{*j_{best}}}{{.2}.N\quad.{h_{p,s}\left( {k - W - {l.N} + {n.N} - u} \right)}}}}}} + {\sum\limits_{n = 0}^{L - 1}\quad{c_{n}^{*_{j_{best}}}.\alpha_{k + {n.\quad N}}}}}$Note also that the weighting code MC^(j) ^(best) ={c₀ ^(j) ^(best) , . .. ,c_(L−1) ^(j) ^(best) } can not be deduced directly from said at leastone output stream z_(s,m)(k_(best)) In that case, the first componentsc₀ ^(j) of each weighting code are all identical avoiding thus at thereceiver any phase ambiguity due to the channel. The information,represented by each weighting code, is then carried by each of (L−1)components c_(q) ^(j)={1, . . . ,L−1} following the first component c₀^(j) and each component c_(q) ^(j)={1, . . . ,L−1} is differentiallyencoded from its first component. A soft estimate ĉ_(q) ^(j) of each of(L−1) components c_(q) ^(j) is then obtained from (L−1) peaksr_(s,m)(k_(best)+N), . . . ,r_(s,m)(k_(best)+(L−1).N) of at least onesingle resulting streams r_(s,m) _(best) (k) related each to the outputstream z_(s,m) _(best) (k) obtained from an antenna RANT_(s) followingthe maximisation of the decision value described later.

According to a first embodiment to retrieve the weighting code MC^(j)^(best) ={c₀ ^(j) ^(best) , . . . ,c_(L−1) ^(j) ^(best) }, the softestimate ĉ_(q) ^(j) ^(best) is given by:${\hat{c}}_{q}^{j_{best}} = \frac{\sum\limits_{s = 1}^{NANT}\quad\left\lbrack {{r_{s,m_{best}}^{*}\left( k_{best} \right)}.{r_{s,m_{best}}\left( {k_{best} + {q.N}} \right)}} \right.}{\sqrt{\sum\limits_{s = 1}^{NANT}\quad{{r_{s,m_{best}}\left( k_{best} \right)}}^{2}}}$

where r_(s,m) _(best) *(k_(best)) is the complex conjugate of the firstpeak and |r_(s,m) _(best) (k_(best))| is the module of the first peak.Note that according to the above equation, the phase rotation andamplitude of the peak r_(s,m) _(best) (k_(best)+q.N) are correctedbefore retrieving the component ĉ_(q) ^(j) ^(best) of the weightingcode.

According to a second embodiment to retrieve the weighting code MC^(j)^(best) ={c₀ ^(j) ^(best) , . . . ,c_(L−1) ^(j) ^(best) }, the softestimate ĉ_(q) ^(j) ^(best) is given by:${\hat{c}}_{q}^{j_{best}} = \frac{\sum\limits_{s = 1}^{NANT}\left\lbrack \frac{\sqrt{G_{s}}.{r_{s,m_{best}}^{*}\left( k_{best} \right)}.{r_{s,m_{best}}\left( {k_{best} + {q.N}} \right)}}{{r_{s,m_{best}}\left( k_{best} \right)}} \right\rbrack}{\sqrt{\sum\limits_{s = 1}^{NANT}\quad G_{s}}\quad}$${{with}\quad G_{s}} = {\frac{\sum\limits_{\ell = 0}^{L - 1}\quad{{r_{s,m_{best}}\left( {k_{best} + {\left( {\ell - 1} \right).N}} \right.}^{2}}}{L}.}$

Note that according to this above equation, the phase rotation andamplitude of the peak r_(s,m) _(best) (k_(best)+q.N) are also correctedbefore retrieving the component ĉ_(q) ^(j) _(best) of the weightingcode.

Each component of the weighting code carried by the receivedsynchronisation sequence is thus retrieved by combining resultingstreams obtained from each antenna RANT_(s). This is a general approachwhich includes the case where a single antenna in considered(N_(ANT)=1).

According to a first embodiment of the time instant determination andweighting code retrieval step 500, at a time instant k, a decision valueis computed for each output streams z_(s,m) ^(j)(k), respectivelyz_(s,m)(k), obtained at the output of the output stream computation step400 if one of the two first embodiments, respectively the thirdembodiment, of the output stream computation step 400 is used.

When an output stream z_(s,m) ^(j)(k) is obtained from the output streamcomputation step 400, according to a first embodiment of the decisionvalue computation, the decision value S_(s,m) ^(j)(k) computed at thetime instant k for said output stream z_(s,m) ^(j)(k) is given byS _(s,m) ^(j)(k)=|z _(s,m) ^(j)(k)|²

and according to a second embodiment, the decision value S_(s,m) ^(j)(k)at the time instant k for said output stream z_(s,m) ^(j)(k) is computedby a correlation merit factor defined by${S_{s,m}^{j}(k)} = \frac{{{z_{s,m}^{j}(k)}}^{2}}{\frac{1}{K_{1} + K_{2}}\left\lbrack {{\sum\limits_{l = 1}^{K_{1}}\quad{{z_{s,m}^{j}\left( {k - l} \right)}}^{2}} + {\sum\limits_{l = 1}^{K_{2}}\quad{{z_{s,m}^{j}\left( {k + l} \right)}}^{2}}} \right\rbrack}$

where |z_(s,m) ^(j)(k)|² is the energy at the time instant k of theoutput stream z_(s,m) ^(j)(k) and$\frac{1}{K_{1} + K_{2}}\left\lbrack {{\sum\limits_{l = 1}^{K_{1}}\quad{{z_{s,m}^{j}\left( {k - l} \right)}}^{2}} + {\sum\limits_{l = 1}^{K_{2}}\quad{{z_{s,m}^{j}\left( {k + l} \right)}}^{2}}} \right\rbrack$is the energy of the output stream z_(s,m) ^(j)(k) averaged on the twotime intervals of size K₁ and K₂ defined respectively before and afterthe time instant k.

The maximisation of the decision value is then given by$\left( {k_{best},j_{best},m_{best},s_{best}} \right) = {\underset{k,m,s,j}{argmax}\left( {S_{s,m}^{j}(k)} \right)}$

independently of the embodiment of the decision value computation.

By selecting the output stream z_(s) _(best) _(,m) _(best) ^(j) ^(best)(k) which maximises the decision value S_(s,m) ^(j)(k) at the timeinstant k_(best), a coarse estimation of the frequency drift is alsoobtained by the possible frequency offset value Δf_(m) _(best) and theweighting code MC^(j) ^(best) related to said output stream z_(s)_(best) _(,m) _(best) ^(j) ^(best) (k) is deduced by the receiver RCD asabove-explained.

FIG. 12 represents a chronogram which depicts the computation of thedecision value S_(s,m) ^(j)(k), according to the second embodiment, atthe time instant (k_(best)−1) and at the time instant k_(best) for anoutput stream z_(s,m) ^(j)(k). The decision value S_(s,m) ^(j)(k) iscomputed at a time instant k from the first time interval defined overK₁ times the duration separating two successive pulses of the outputstream z_(s,m) ^(j)(k). Said first time interval ends at the timeinstant (k−1). The decision value S_(s,m) ^(j)(k) is also computed fromthe second time interval defined over K₂ times the duration separatingtwo successive pulses of the output stream z_(m) ^(j)(k_(r)). Saidsecond time interval starts at the time instant (k+1).

The example shows that the decision value S_(s,m) ^(j)(k) is maximal atthe time instant k_(best), i.e. when the decision value is computed atthe time instant at which the single peak appears in the output streamz_(s,m) ^(j)(k).

When an output stream z_(s,m)(k) is obtained from the output streamcomputation step 400, according to an embodiment of the decision valuecomputation, the decision value S_(s,m)(k) computed at the time instantk for said output stream z_(s,m)(k) is given byS _(s,m)(k)=|Z _(s,m)(k)|²

and according to another embodiment, the decision value S_(s,m)(k) atthe time instant k for said output stream z_(s,m)(k) is computed by acorrelation merit factor defined by${S_{s,m}(k)} = \frac{{{z_{s,m}(k)}}^{2}}{\frac{1}{K_{1} + K_{2}}\left\lbrack {{\sum\limits_{l = 1}^{K_{1}}\quad{{z_{s,m}\left( {k - l} \right)}}^{2}} + {\sum\limits_{l = 1}^{K_{2}}\quad{{z_{s,m}\left( {k + l} \right)}}^{2}}} \right\rbrack}$

where |z_(s,m)(k)|² is the energy at the time instant k of the outputstream z_(s,m)(k) and$\frac{1}{K_{1} + K_{2}}\left\lbrack {{\sum\limits_{l = 1}^{K_{1}}\quad{{z_{s,m}\left( {k - l} \right)}}^{2}} + {\sum\limits_{l = 1}^{K_{2}}\quad{{z_{s,m}\left( {k + l} \right)}}^{2}}} \right\rbrack$is the energy of the output stream z_(s,m)(k)averaged on the two timeintervals of size K₁ and K₂ defined respectively before and after thetime instant k.

The maximisation of the decision value is then given by$\left( {k_{best},m_{best}} \right) = {\underset{k,m}{\arg\quad\max}\left( {S_{s,m}(k)} \right)}$independently of the embodiment of the decision value computation.

By selecting the output streams z_(s,m) _(best) (k) which maximise thedecision value S_(s,m)(k) at time instant k_(best), a coarse estimationof the frequency drift is obtained by the possible frequency offsetvalue Δf_(m) _(best) .

According to a second embodiment of the time instant determination andweighting code retrieval step 500, at a time instant k, a decision valueis computed for at least one combination of either output streamsz_(s,m) ^(j)(k), respectively z_(s,m)(k), obtained at the output of theoutput stream computation step 400 if one of the two first embodiments,respectively the third embodiment, of the output stream computation step400 is used.

When one of the two first embodiments of the output stream computationstep 400 is used, a combination of output streams z_(s,m) ^(j)(k) isrelated to either one of I weighting codes MC^(j) or one of M frequencyoffsets Δf_(m) or one of M frequency offsets Δf_(m) and one of Iweighting codes MC^(j).

In case M=1, when a combination of output streams z_(s,1) ^(j)(k)isrelated to a weighting code MC^(i), only the output streams z_(s,1)^(j=i)(k) related to this weighting code are considered from eachantenna RANT_(s). When a combination of output streams z_(s,m) ^(j)(k)is related to both a frequency offset Δf_(μ) and a weighting codeMC^(i), only one output stream z_(s,m=μ) ^(j=i)(k) is considered fromeach antenna RANT_(s).

When the third embodiment of the output stream computation step 400 isused, a combination of output streams z_(s,m)(k) is related to one of Mfrequency offsets Δf_(m).

When a combination of output streams z_(s,m) ^(j)(k) obtained from theoutput stream computation step 400 is considered, according to anembodiment of the decision value computation, the decision value S_(m)^(j)(k) computed at the time instant k for said combination is given by${S_{m}^{j}(k)} = {\sum\limits_{s = 1}^{NANT}{{z_{s,m}^{j}(k)}}^{2}}$

and according to another embodiment, the decision value S_(m) ^(j)(k) atthe time instant k for said combination is computed by a correlationmerit factor defined by${S_{m}^{j}(k)} = \frac{\sum\limits_{s = 1}^{NANT}{{z_{s,m}^{j}(k)}}^{2}}{\frac{1}{N_{ANT}\left( {K_{1} + K_{2}} \right)}{\sum\limits_{s = 1}^{NANT}\left\lbrack {{\sum\limits_{l = 1}^{K_{1}}{{z_{s,m}^{j}\left( {k - l} \right)}}^{2}} + {\sum\limits_{l = 1}^{K_{2}}{{z_{s,m}^{j}\left( {k + l} \right)}}^{2}}} \right\rbrack}}$

The maximisation of the decision value is then given by$\left( {k_{best},j_{best},m_{best}} \right) = {\underset{k,m,j}{\arg\quad\max}\left( {S_{m}^{j}(k)} \right)}$

independently of the embodiment of the decision value computation.

By selecting the set of output streams z_(s,m) _(best) ^(j) ^(best) (k)which maximises the decision value S_(m) ^(j)(k) at the time instantk_(best), an estimation of the frequency drift is obtained by thepossible frequency offset value Δf_(m) _(best) . The weighting codeMC^(j) ^(best) is also deduced from said set of output streams z_(s,m)_(best) ^(j) ^(best) (k) by the receiver RCD as above-explained.

When a combination of output streams z_(s,m)(k) obtained from the outputstream computation step 400 is considered, according to an embodiment ofthe decision value computation, the decision value S_(m)(k) computed atthe time instant k for said combination is given by${S_{m}(k)} = {\sum\limits_{s = 1}^{NANT}{{z_{s,m}(k)}}^{2}}$

and according to another embodiment, the decision value S_(m)(k) at thetime instant k for said combination is computed by a correlation meritfactor defined by${S_{m}(k)} = \frac{\sum\limits_{s = 1}^{NANT}{{z_{s,m}(k)}}^{2}}{\frac{1}{N_{ANT}\left( {K_{1} + K_{2}} \right)}{\sum\limits_{s = 1}^{NANT}\left\lbrack {{\sum\limits_{l = 1}^{K_{1}}{{z_{s,m}\left( {k - l} \right)}}^{2}} + {\sum\limits_{l = 1}^{K_{2}}{{z_{s,m}\left( {k + l} \right)}}^{2}}} \right\rbrack}}$

The maximisation of the decision value is then given by$\left( {k_{best},m_{best}} \right) = {\underset{k,m}{\arg\quad\max}\left( {S_{m}(k)} \right)}$

independently of the embodiment of the decision value computation.

By selecting the set of output streams z_(s,m) _(best) (k) whichmaximises the decision value S_(m)(k) at the time instant k_(best), anestimation of the frequency drift is obtained by the possible frequencyoffset value Δf_(m) _(best) . As explained in the opening paragraph, afine frequency correction of the received signal frame may be required.The phase of the received signal evolves linearly in time according to aslope value Δφ. The phase of the received signal may then be correctedby the receiver RCD, when the slope value Δφ is estimated.

The classical way to perform an estimation of such a slope value Δφ isto use two or more repetitions of a same sequence of pulses included inthe synchronisation sequence y_(s)(k), and to compute the phase rotationbetween the repetitions (T. M. Schmidl, D. C. Cox, “Robust frequency andtiming synchronization for OFDM,” IEEE Transactions on Communications,vol. 45, pp. 1613-1621, no. 12, December 1997). For instance, therespective phases encountered by the first pulse of two consecutivesequences of pulses are e^(j2π(f) ⁰ ^(+Δf) ^(′) ^()t) ⁰ ^(+φ) ⁰ ande^(j2π(f) ⁰ ^(+Δf) ^(′) ^()(t) ⁰ ^(+Δt)+φ) ⁰ , where Δt is the time gapbetween the first pulse of the two sequences of pulses. In base band,the phase rotation between the two consecutive sequences of pulses ishence Δφ=2πΔf^(′)Δt+2zπ, where Δf^(′) the frequency offset related tothe phase rotation.

To avoid any phase ambiguity it is necessary to respect −π<Δφ<π whichprovides the frequency offset estimation range${- \frac{1}{2\Delta\quad t}} < {\Delta\quad f^{\prime}} < \frac{1}{2\Delta\quad t}$or equivalently${- \frac{f_{s}}{V}} < {\Delta\quad f^{\prime}} < {\frac{f_{s}}{2V}.}$

Obviously, the higher V, the smaller the frequency range.

The slope Δφ may be then estimated for computing a single estimation ofthis slope when the phase rotation between the first pulses of the twosequences is the same that the phase rotation between the second pulseswhich is the same that the phase rotation between the third pulses, andso on. This occurs when the pulses of the two received sequences remainidentical after having been filtered by the P paths channel$\sum\limits_{p = 0}^{P - 1}{{h_{p,s}(k)} \cdot {\delta\left( {k - p} \right)}}$where δ(k) is the Dirac function.

In the present invention, the pulses of the two sequences from which anestimation of the slope is computed, for example two consecutivecomplementary sequences A, are multiplied by the components of aweighting code. This multiplication involves that the two receivedsequences are not always identical after having been filtered by the Ppaths channel. Consequently, the previous estimation approach shall beadapted by using the perfect autocorrelation property of complementarysequences.

According to an embodiment of the present invention, when the receivedsignal frame is synchronised in time and possibly coarsely corrected infrequency, in the course of the time instant determination and weightingcode retrieval step 500 the phase of each pulse of the received signalframe is corrected by a linearly increasing value the slope of which,{circumflex over (Δ)}φ, is estimated from a weighted average of P slopeestimations {circumflex over (Δ)}φ_(p,s) obtained for each of said atleast one antenna RANT_(s). The estimate of the slope {circumflex over(Δ)}φ is thus obtained by${\hat{\Delta}\quad\varphi} = \frac{\sum\limits_{s = 1}^{NANT}{\sum\limits_{p = 0}^{P - 1}\left\lbrack {{{{\hat{h}}_{p,s}}^{2} \cdot \hat{\Delta}}\quad\varphi_{p,s}} \right\rbrack}}{\sum\limits_{s = 1}^{NANT}{\sum\limits_{p = 0}^{P - 1}{{\hat{h}}_{p,s}}^{2}}}$

where |ĥ_(p,s)|² is an estimate of the squared amplitude of the pathcoefficient p for an antenna RANT_(s) and {circumflex over (Δ)}φ_(p,s)is the estimate of the slope related to said path p.

According to an embodiment of the estimation of a slope {circumflex over(Δ)}φ_(p,s), the estimate {circumflex over (Δ)}φ_(p,s) is obtained by${\hat{\Delta}\quad\varphi_{p,s}} = \frac{\hat{\Delta}\quad\varphi_{p,s}}{N}$

where {circumflex over (Δ)}φ_(p,s) ^(′) is the sequence by sequenceestimation of the slope given by${\hat{\Delta}\quad\varphi_{p,s}} = \frac{\sum\limits_{\ell = 1}^{L - 1}\left\lbrack {{\varphi\left\lbrack {C_{A,p,s}^{\ell} + C_{B,p,s}^{\ell}} \right\rbrack} - {\varphi\left\lbrack {C_{A,p,s}^{\ell - 1} + C_{B,p,s}^{\ell - 1}} \right\rbrack}} \right\rbrack}{L - 1}$

where C_(A,p,s) ^(l) and C_(B,p,s) ^(l) are respectively a first and asecond correlation factors computed from respectively the first and thesecond complementary sequence of rank l of the received synchronisationsequence, and φ(C_(A,p,s) ^(l)+C_(A,p,s) ^(l)) is the phase of the sumof the first and second correlation factors.

The squared amplitude of a path coefficient p for an antenna RANT_(s) isthen estimated by${{\hat{h}}_{p,s}}^{2} = {\frac{1}{L}{\sum\limits_{\ell = 0}^{L - 1}{{C_{A,p,s}^{\ell} + C_{B,p,s}^{\ell}}}^{2}}}$

FIG. 13 shows how two correlation factors C_(A,p,s) ^(l) and C_(B,p,s)^(l) related to a path p are computed.

The first correlation factors C_(A,p,s) ^(l)l∈{0, . . . ,L−1} related toa path p are computed at a time instant k from a N pulses long segmentof a first complementary sequence A by $\begin{matrix}{C_{A,p,s}^{\ell} = {\sum\limits_{k = {W + p}}^{W + p + N - 1}{{y_{s}(k)} \cdot {A(k)}}}} \\{= {{\sum\limits_{k = 0}^{N - 1}{\sum\limits_{p = 0}^{P - 1}{{h_{p,s} \cdot {A\left( {k - p} \right)}}{{\mathbb{e}}^{{j\quad 2\quad\pi\quad\Delta\quad{{f^{\prime}{({k + W})}}/f_{s}}} + {j\quad\varphi_{0}}} \cdot {A(k)}}}}} + {\sum\limits_{k = {W + p}}^{W + p + N - 1}{{n(k)} \cdot {A(k)}}}}}\end{matrix}$

The second correlation factors C_(B,p,s) ^(l)l∈{0, . . . ,L−1} relatedto a path p are computed at a time instant k from a N pulses longsegment of a first complementary sequence B by $\begin{matrix}{C_{B,p,s}^{\ell} = {\sum\limits_{k = {W + p}}^{W + p + N - 1 + u}{{y_{s}(k)} \cdot {B(k)}}}} \\{= {{\sum\limits_{k = 0}^{N - 1}{\sum\limits_{p = 0}^{P - 1}{{h_{p,s} \cdot {B\left( {k - p} \right)}}{{\mathbb{e}}^{{j\quad 2\quad\pi\quad\Delta\quad{{f^{\prime}{({k + W + u})}}/f_{s}}} + {j\quad\varphi_{0}}} \cdot {B(k)}}}}} +}} \\{\sum\limits_{k = {W + p}}^{W + p + N - 1}{{n(k)} \cdot {B(k)}}}\end{matrix}$

By summing the two correlation factors and using the perfectautocorrelation property of complementary sequences given by equation(1), we obtainC _(A,p,s) ^(l) +C _(B,p,s) ^(l) =h _(p) l ^(j2πΔf) ^(′) ^(N(l−1)/f)^(s) ^(+jφ) ⁰

The stream which carries a single peak from which a signal frame issynchronised in time exhibits possibly secondary peaks which are due tobad autocorrelation properties of weighting codes. This drasticallyincreases the detection error probability of the single peak. Accordingto one of the embodiments of the synchronisation method or one of theirvariants, in order to counter fight this problem, the I weighting codesMC^(j) are multiplied by a same scrambling code in the course of thesynchronisation sequence generation step 100, and the I weighting codesMC^(j), used in the course of the single peak detection step 300 arealso multiplied by said scrambling code. Such a scrambling is, forexample the well-known Barker code (“Sequence design for communicationsapplications” P. Fan, M. Darnell, , pp.270-272, Wiley, New York, 1996),)which bounds the norm of said secondary peaks to be N times lower thanthe norm of the main peak.

According to another aspect of the present invention, the signal Tsg(k)is transmitted from multiple antennas TANT_(s). The method forsynchronising according to the present invention is then characterisedin that the transmitter TRD transmits the same signal Tsg(k) from eachof said antenna TANT_(s) with minor different time delays.

According to one of its hardware-oriented aspects, the present inventionrelates to the transmitter TRD and the receiver RCD of thetelecommunication system SYST.

FIG. 14 represents a schema of the transmitter TRD intended to executethe synchronisation sequence generation step 100 and the sequenceweighting step 200 according to the present invention.

The transmitter TRD is intended to generate a synchronisation sequencex^(i)(k) given by equation (2) and to transmit to the receiver RCD asignal frame which includes said generated synchronisation sequence. Forgenerating the synchronisation sequence x^(i)(k), the transmitter TRDincludes a pulse generator SGM for generating a pair of complementarysequences A and B, a burst creator BC for creating a first burst of Lsequences A and a second burst of L sequences B from said pair ofcomplementary sequences (A,B), and a multiplier WGM for multiplying thepulses of each complementary sequence A of rank q of the first burst ofL sequences by the component of rank q of a L components long weightingcode MC^(i), and for multiplying the pulses of each sequence of rank qof the second burst of L sequences by the same component of rank q ofsaid weighting code MC^(i).

According to an embodiment of the burst creator BC, at least one guardinterval, defined by a W bits long cyclic extension of either the firstor second complementary sequence is included in the synchronisationsequence x^(i)(k). Said at least one guard interval is located either atthe beginning or at the end of the first burst of L sequences when saidat least one guard interval is defined from the first complementarysequence A, and said at least one guard interval is located either atthe beginning or at the end of the second burst of L sequences when saidat least one guard interval is defined from the second complementarysequence B.

According to an embodiment of the multiplier WGM, the weighting codeMC^(i) belonging to a set of I weighting codes, the set of I weightingcodes is a set of orthogonal codes from which a fast transform may beobtained. For example, each weighting code is a Hadamard code or aFourier code.

According to a variant of this embodiment, the weighting codes MC^(i)are multiplied by a same scrambling code which is for example a Barkercode.

The transmitter TRD is equipped by at least one antenna TANT from whichthe signal Tsg(k) is transmitted. When the transmitter TRD is equippedby multiple antennas TANT according to an embodiment of the presentinvention, the same signal Tsg(k) is transmitted from each of saidantenna with minor different time delays. Note that the transmission onmultiple antennas TANT does not have any impact on the processingsexecuted by the receiver RCD which remain identical to thoseabove-described.

FIG. 15 represents a schema of the receiver RCD intended to synchronisein time and possibly correct in frequency a received signal frameaccording to the present invention.

The receiver RCD, which is equipped by at least one antennas RANT_(s)intended each to receive a signal frame, is characterised in that itincludes for each of said at least one antenna RANT_(s), means RSM forobtaining at least one resulting stream from the signal which is beingreceived by the antenna RANT_(s).

The means RSM includes

-   -   at least one time-shifter TSV for computing a time-delayed        version of the received signal,    -   at least one first correlator FCS1 for correlating the        time-delayed version of the received signal with a replica of        the first complementary sequence A of the pair of complementary        sequences (A,B),    -   at least one second correlator FCS2 for correlating the        time-delayed version of the received signal with a replica of        the first complementary sequence B of the pair of complementary        sequences (A,B), and    -   at least one correlation stream combiner SA for forming a        resulting stream from the first correlation stream and the        second correlation stream.

For example, if Golay sequences are used the correlators FCS1 and FCS2are replaced by a well-known Extended Golay Correlator.

Moreover, the receiver RCD includes a resulting stream combiner RSC forobtaining at least one output stream for each of said at least oneresulting stream, means DV for computing at least one decision value ata time instant from said at least one output stream or at least onecombination of said output streams and means DVM for maximising said atleast one decision value.

According to an embodiment of the receiver RCD, it includes also meansCFM1 for correcting in frequency the phase of pulses of a secondcorrelation stream related to a same complementary sequence by aconstant value related to a predefined frequency offset value.

According to another embodiment of the receiver RCD, it includes meansCFM2 for correcting the phase of each pulse of the received signal by alinearly increasing value the slope of which is related to a predefinedfrequency offset value.

According to a first embodiment of the means DV, each decision valuecomputed at a time instant related to an output stream, respectively acombination of output streams, is the squared norm of said outputstream, respectively the combined output stream, evaluated at said timeinstant.

According to a second embodiment of the means DV, each decision valuecomputed at a time instant related to an output stream, respectively acombination of output streams, is a correlation merit factor asabove-described

According to an embodiment of the resulting stream combiner RSC,depicted in FIG. 16 a, the resulting stream combiner RSC is a bunch ofcorrelators COj which are each intended to correlate at a given timeinstant a resulting stream with a comb of pulses related to a weightingcode as above-explained.

According to another embodiment of the resulting stream combiner RSC,depicted in FIG. 16 b, the weighting code belonging to a set of Iweighting codes which is a set of orthogonal codes from which a fasttransform may be obtained, the resulting stream combiner RSC includesmeans for processing L pulses of a resulting stream separated to eachother by N positions by said fast transform.

According to another embodiment of the resulting stream combiner RSC,depicted in FIG. 16 c, the resulting stream combiner RSC includes meansfor computing at a time instant the square root of the sum of the energyof L pulses separated by N positions from each other of said resultingstream.

The receiver RCD includes, according to this embodiment, a weightingcode retriever WCR which includes means for obtaining a soft estimate ofa component of rank q of the weighting code carried by the receivedsignal frame from the product of the sum of the phase rotation andamplitude corrected peaks of rank q of said at least one singleresulting stream by a weighting value. The phase rotation and amplitudeof peaks of rank q are corrected as above-explained.

According to another embodiment of the receiver RCD, the receiver RCDincludes means for correcting the phase of each pulse of a receivedsignal frame by a linearly increasing value the slope of which isobtained from a weighted average of at least P slope estimationsobtained for each of said at least one antennas. The P slope estimationsare obtained as above-explained.

1) Method for synchronising a signal frame transmitted by a transmitterof a telecommunication system to a receiver adapted to synchronise saidsignal frame from a synchronisation sequence included in said signalframe, characterised in that it includes: a synchronisation sequencegeneration step (100) intended to be executed by the transmitter in thecourse of which the synchronisation sequence (x^(i)(k)) is formed by theconcatenation of a first and a second bursts of L complementarysequences, said first burst of L sequences being obtained by theconcatenation of L times a first N pulses complementary sequence (A) ofa pair of complementary sequences ((A,B)), said second burst of Lsequences being obtained by the concatenation of L times a second Npulses complementary sequence (B) of said pair of complementarysequences, and a sequence weighting step (200) intended to be executedby the transmitter in the course of which the pulses of eachcomplementary sequence (A) of rank q of the first burst of L sequencesare multiplied by the component of rank q of a L components longweighting code (MC^(i)) belonging to a set of I weighting codes known bythe receiver (RCD) beforehand, and the pulses of each sequence of rank qof the second burst of L sequences are multiplied by the same componentof rank q of said weighting code (MC^(i)). 2) Method for synchronisingas claimed in claim 1, characterised in that in the course of thesynchronisation sequence generation step (100) at least one guardinterval, defined by a W bits long cyclic extension of either the firstor second complementary sequence is included in the synchronisationsequence (x^(i)(k)), said at least one guard interval being locatedeither at the beginning or at the end of the first burst of L sequenceswhen said at least one guard interval is defined from the firstcomplementary sequence (A), and said at least one guard interval beinglocated either at the beginning or at the end of the second burst of Lsequences when said at least one guard interval is defined from thesecond complementary sequence (B). 3) Method for synchronising asclaimed in claim 2, characterised in that two guard intervals (Δ_(A)^(Le), Δ_(A) ^(Ri)) are defined by W bits long cyclic extensions of thefirst complementary sequence (A), one of said guard interval (Δ_(A)^(Le)) being located at the beginning of the first burst of L sequencesand the other one (Δ_(A) ^(Ri)) being located at the end of the firstburst of L sequences, and two guard intervals (Δ_(B) ^(Le), Δ_(B) ^(Ri))are defined by W bits long cyclic extensions of the second complementarysequence (B), one of said guard interval (Δ_(B) ^(Le)) being located atthe beginning of the second burst of L sequences and the other one(Δ_(B) ^(Ri)) being located at the end of the second burst of Lsequences. 4) Method for synchronising as claimed in one of claims 1 to3, characterised in that the set of I weighting codes is a set oforthogonal codes from which a fast transform is obtained. 5) Method forsynchronising as claimed in claim 4, characterised in that eachweighting code is a Hadamard code. 6) Method for synchronising asclaimed in claim 4, characterised in that each weighting code is aFourier code. 7) Method for synchronising as claimed in one of claims 4to 6, characterised in that the weighting codes are multiplied by a samescrambling code. 8) Method for synchronising as claimed in claim 7,characterised in that the scrambling code is a Barker code. 9) Methodfor synchronising as claimed in one of claims 1 to 8, the receiver beingequipped by at least one antenna intended each to receive said signalframe, characterised in that it further includes, for each of said atleast one antenna (RANT_(s)), a resulting stream computation step (300_(s)) in the course of which at least one resulting stream is obtainedfrom the signal which is being received by the antenna (RANT_(s)), saidresulting stream computation step (300 _(s)) includes at least onetime-shifting sub-step (301 _(s)) in the course of each a time-delayedversion of the received signal is computed, at least one first sequencecorrelation sub-step (302 _(s)) in the course of each a firstcorrelation stream is computed by correlating the time-delayed versionof the received signal with a replica of the first complementarysequence (A) of the pair of complementary sequences (A,B), at least onesecond sequence correlation sub-step (303 _(s)) in the course of each asecond correlation stream is computed by correlating the received signalwith a replica of the second complementary sequence (B) of the pair ofcomplementary sequences (A,B), at least one stream summation sub-step(304 _(s)) in the course of each a resulting stream is formed from thefirst correlation stream and the second correlation stream, an outputstream computation step (400) in the course of which at least one outputstream is obtained for each of said at least one resulting stream, and atime instant determination and weighting code retrieval step (500) inthe course of which at least one decision value at a time instant iscomputed from said at least one output stream, a time instant(k_(best)), at which the signal frame which is being received is saidsynchronised in time, is then obtained by maximising said at least onedecision value, and the weighting code (MC^(j) ^(best) ) carried by thereceived synchronisation sequence is retrieved from L peaks separated byN positions from each other of at least one of said resulting streamsobtained from said maximum decision value, the first of said L peaks ofeach of said at least one resulting streams being located at the timeinstant (k_(best)) at which the signal frame is said synchronised intime. 10) Method for synchronising as claimed in claim 9, characterisedin that said at least one resulting stream computation step (300 _(s))includes a single time-shifting sub-step (301 _(s)) in the course ofwhich a time-delayed version of the received signal is computed, asingle first sequence correlation sub-step (302 _(s)) in the course ofwhich a first correlation stream (y_(A,s)(k)) is computed by correlatingthe time-delayed version of the received signal with a replica of thefirst complementary sequence (A) of the pair of complementary sequences(A,B), a single second sequence correlation sub-step (303 _(s)) in thecourse of which a second correlation stream (y_(B,s)(k)) is computed bycorrelating the received signal with a replica of the secondcomplementary sequence (B) of the pair of complementary sequences (A,B),and a single stream summation sub-step (304 _(s)) in the course of whicha resulting stream (r_(s)(k)) is formed by summing said firstcorrelation stream (y_(A,s)(k)) and said second correlation stream(y_(B,s)(k)). 11) Method for synchronising as claimed in claim 9, Mfrequency offset values being predefined by the receiver from a range ofpossible frequency drifts, characterised in that said at least oneresulting stream computation step (300 _(s)) includes a singletime-shifting sub-step (301 _(s)) in the course of which a time-delayedversion of the received signal is computed, a single first sequencecorrelation sub-step (302 _(s)) in the course of which a firstcorrelation stream (y_(A,s)(k)) is computed by correlating thetime-delayed version of the received signal with a replica of the firstcomplementary sequence (A) of the pair of complementary sequences (A,B),a single second sequence correlation sub-step (303 _(s)) in the courseof which a second correlation stream (y_(B,s)(k)) is computed bycorrelating the received signal with a replica of the secondcomplementary sequence (B) of the pair of complementary sequences (A,B),M coarse sequence by sequence frequency correction sub-steps (305_(s,m)) in the course of each the phase of pulses of said secondcorrelation stream (y_(B,s)(k)) related to a same complementary sequenceis corrected by a constant value (Δφ_(m)) related to one of said Mfrequency offset values, and M stream summation sub-steps (304 _(s,m))in the course of each a resulting stream (r_(s,m)(k)) is formed bysumming the first correlation stream (y_(A,s)(k)) and one of said Mcorrected second correlation streams (y_(B,s,m)(k)). 12) Method forsynchronising as claimed in claim 9, M frequency offset values beingpredefined by the receiver from a range of possible frequency drifts,characterised in that said at least one resulting stream computationstep (300 _(s)) includes M coarse pulse by pulse frequency correctionsub-steps (306 _(s,m)) in the course of each the phase of each pulse ofthe received signal is corrected by a linearly increasing value theslope of which (Δφ_(m)) is related to one of said M frequency offsetvalues, M time-shifting sub-steps (301 _(s,m)) in the course of each atime-delayed version of one of said M received and phase correctedsignals (y_(s,m)(k) ) is computed, M first sequence correlationsub-steps (302 _(s,m)) in the course of each a first correlation stream(y_(A,s,m)(k)) is computed by correlating the time-delayed version ofone of said M received and phase corrected signals with a replica of thefirst complementary sequence (A) of the pair of complementary sequences(A,B), M second sequence correlation sub-steps (303 _(s,m)) in thecourse of each a second correlation stream (y_(B,s,m)(k)) is computed bycorrelating one of said M received and phase corrected signals(y_(s,m)(k)) with a replica of the second complementary sequence (B) ofthe pair of complementary sequences (A,B), and M stream summationsub-steps (304 _(s,m)) in the course of each a resulting stream(r_(s,m)(k)) is formed from one of said M first correlation streams(y_(A,s,m)(k)) and one of said M second correlation streams(y_(B,s,m)(k)) related to the same received and phase corrected signal(y_(s,m)(k)). 13) Method for synchronising as claimed in one of claims 9to 12, characterised in that in the course of the time instantdetermination and weighting code retrieval step (500) at a time instanta decision value is computed for each of said at least one output streamobtained from each of said at least one antenna. 14) Method forsynchronising as claimed in one of claims 9 to 12, characterised in thatin the course of the time instant determination and weighting coderetrieval step (500) at a time instant a decision value is computed forat least one combination of said at least one output stream, each ofsaid at least one combination, which is related to either one of said Iweighting codes or one of said M frequency offsets and one of said Iweighting codes, is defined by the square root of the sum of squaredmodules of output streams obtained from said at least one antenna whichare related to either a same weighting code or the same weighting codeand the same frequency value. 15) Method for synchronising as claimed inone of claims to 13 to 14, characterised in that in the course of thetime instant determination and weighting code retrieval step (500) eachdecision value computed at a time instant related to an output stream,respectively a combination of output streams, is the squared norm ofsaid output stream, respectively the combined output stream, evaluatedat said time instant. 16) Method for synchronising as claimed in one ofclaims 13 to 14, characterised in that in the course of the time instantdetermination and weighting code retrieval step (500) each decisionvalue computed at a time instant related to an output stream,respectively a combination of output streams, is a correlation meritfactor defined by the ratio of the energy at said time instant of saidoutput stream, respectively said combination of output streams, dividedby the energy of said output stream, respectively said combination ofoutput streams, averaged on two time intervals defined respectivelybefore and after said time instant. 17) Method for synchronising asclaimed in one of claims 9 to 16, characterised in that in the course ofthe output stream computation step (400), I output streams (z_(s,m)^(j)(k)) are obtained for each of said at least one resulting stream,each of said I output streams related to one of said at least oneresulting stream being obtained by correlating at a given time instantsaid resulting stream with a comb of pulses (cp^(j)) related to one ofsaid I weighting codes, each pulse of a comb related to a weighting codebeing separated to each other by N positions and each pulse of said combbeing weighted by a component of said weighting code. 18) Method forsynchronising as claimed in one of claims 9 to 16, the set of Iweighting codes being a set of orthogonal codes from which a fasttransform is obtained, characterised in that in the course of the outputstream computation step (400), I output streams (z_(s,m) ^(j)(k)) areobtained for each of said at least one resulting stream, each of said Ioutput streams related to each of said at least one resulting streambeing obtained by processing L pulses of said resulting stream separatedto each other by N positions with said fast transform. 19) Method forsynchronising as claimed in one of claims 9 to 16, characterised in thatin the course of the output stream computation step (400), a singleoutput stream (z_(s,m)(k)) is obtained for each of said at least oneresulting stream by computing at a time instant the squared root of thesum of the energy of L pulses separated by N positions from each otherof said resulting stream. 20) Method for synchronising as claimed inclaim 19, each of (L−1) components following the first component of eachof said I weighting codes being differentially encoded from its firstcomponent, the signal frame being possibly corrected in frequency, Lpeaks separated by N positions from each other of a single resultingstream (r_(s,m) _(best) (k)) obtained from said maximal decision valueand related to each of said at least one antenna being considered, thefirst of said L peaks being located at the time instant (k_(best)) atwhich the signal frame is synchronised in time, characterised in that asoft estimate (ĉ_(q) ^(j) ^(best) ) of a component of rank q of theweighting code (MC^(j) ^(best) ) carried by the received signal frame isobtained from the product of the sum of the phase rotation and amplitudecorrected peaks of rank q (r_(s,m) _(best) (k_(best)+q.N) ) of said atleast one single resulting stream by a weighting value. 21) Method forsynchronising as claimed in claim 20, characterised in that the phaserotation and amplitude of said at least one peak of rank q are correctedby multiplying said at least one peak of rank q by the complex conjugateof the peak preceding said (L−1) peaks (r_(s,m) _(best) *(k_(best))),and said weighting value is the square root of the sum of the square ofthe module of peaks preceding the (L−1) peaks of each of said at leastone single resulting stream. 22) Method for synchronising as claimed inclaim 20, characterised in that the phase rotation and amplitude of saidat least one peak of rank q are corrected by multiplying said at leastone peak of rank q by the product of the complex conjugate of the peakpreceding said (L−1) peaks divided by its module by the square root ofthe average energy of said L peaks, and said weighting value is thesquare root of sum of the average energies of said L peaks obtained foreach of said at least one single resulting stream. 23) Method forsynchronising as claimed in one of claims 1 to 22, the signal framereceived by each of said at least one antenna being transmitted on amultipath channel having P consecutive paths and being synchronised intime and possibly coarsely corrected in frequency, characterised in thatin the time instant determination and weighting code retrieval step(500) the phase of each pulse of a received signal frame is corrected bya linearly increasing value the slope of which ({circumflex over (Δ)}φ)is obtained from a weighted average of at least P slope estimations({circumflex over (Δ)}φ_(p,s)) obtained for each of said at least oneantennas, each of said P slope estimations being weighted by an estimateof the squared amplitude of one of said P path coefficients, theweighting value of said average being the sum of the squared amplitudeof said P path coefficients. 24) Method for synchronising as claimed inclaim 23, characterised in that each slope estimation related to a pathcoefficient is obtained by the ratio of a sequence by sequence slopeestimation ({circumflex over (Δ)}φ_(p,s) ^(′)) related to said pathcoefficient over N, said sequence by sequence slope estimation beingdefined by the average of (L−1) differences between the phase of the sumof a first and a second correlation factors (C_(A,p,s) ^(l)+C_(B,p,s)^(l)) related to said path coefficient and computed at a first timeinstant (l) on a segment of respectively a first and secondcomplementary sequence of the received synchronisation sequence and thephase of the sum of a first and a second correlation factors (C_(A,p,s)^(l−1)+C_(B,p,s) ^(l−1)) related to said path coefficient and computedat a second time instant (l−1) on a segment of respectively a first andsecond complementary sequence of the received synchronisation sequence.25) Method for synchronising as claimed in one of claims 1 to 24, thesignal frame being transmitted from multiple antennas, characterised inthat the transmitter transmits the same signal frame from each of saidantenna with minor different time delays. 26) Method for synchronisingas claimed in on of claims 1 to 25, characterised in that thecomplementary sequences are Golay sequences. 27) Transmitter of atelecommunication system intended to transmit a signal frame to areceiver adapted to synchronise said signal frame from a synchronisationsequence included in said signal frame, characterised in that itincludes: a pulse generator (SGM) for generating a pair of complementarysequences (A,B), a burst creator (BC) for creating a first burst of Lsequences (A) and a second burst of L sequences (B) from said pair ofcomplementary sequences (A,B), a multiplier (WGM) for multiplying thepulses of each complementary sequence (A) of rank q of the first burstof L sequences by the component of rank q of a L components longweighting code (MC^(i)), and for multiplying the pulses of each sequence(B) of rank q of the second burst of L sequences by the same componentof rank q of said weighting code (MC^(i)). 28) Transmitter as claimed inclaim 27, characterised in that at least one guard interval, defined bya W bits long cyclic extension of either the first or secondcomplementary sequence is included in the synchronisation sequence(x^(i)(k)), said at least one guard interval is located either at thebeginning or at the end of the first burst of L sequences when said atleast one guard interval is defined from the first complementarysequence (A), and said at least one guard interval is located either atthe beginning or at the end of the second burst of L sequences when saidat least one guard interval is defined from the second complementarysequence (B). 29) Transmitter as claimed in one of claims 27 to 28,characterised in that the set of I weighting codes is a set oforthogonal codes from which a fast transform may be obtained. 30)Transmitter as claimed in claim 29, characterised in that each weightingcode is a Hadamard code. 31) Transmitter as claimed in claim 29,characterised in that each weighting code is a Fourier code. 32)Transmitter as claimed in one of claims 29 to 31, characterised in thatthe weighting codes are multiplied by a same scrambling code. 33)Transmitter as claimed in one of claims 27 to 32, the transmitter beingequipped by multiple antennas from which the signal is transmitted,characterised in that the same signal is transmitted from each of saidantenna with minor different time delays 34) Receiver of atelecommunication system intended to synchronise in time and possibly tocorrect in frequency a received signal frame, the receiver beingequipped by at least one antenna intended each to receive said signalframe, characterised in that it includes for each of said at least oneantenna, means (RSM) for obtaining at least one resulting stream fromthe signal which is being received by the antenna (s), said means (RSM)include at least one time-shifter (TSV) for computing a time-delayedversion of the received signal, at least one first correlator (FCS1) forcorrelating the time-delayed version of the received signal with areplica of the first complementary sequence (A) of the pair ofcomplementary sequences ((A,B)), at least one second correlator (FCS2)for correlating the time-delayed version of the received signal with areplica of the first complementary sequence (A) of the pair ofcomplementary sequences ((A,B)), at least one correlation streamcombiner (SA) for forming a resulting stream from the first correlationstream and the second correlation stream. a resulting stream combiner(RSC) for obtaining at least one output stream for each of said at leastone resulting stream, means (DV) for computing at least one decisionvalue at a time instant from said at least one output streams or atleast one combination of said output streams, and means (DVM) formaximising said at least one decision value. 35) Receiver as claimed inclaim 34, characterised in that said at least one first and secondcorrelators (FCS1,FCS2) are an Extended Golay Correlator. 36) Receiveras claimed in one of claims 34 or 35, characterised in that it includesalso means (CFM1) for correcting in frequency the phase of pulses of asecond correlation stream related to a same complementary sequence by aconstant value related to a predefined frequency offset value. 37)Receiver as claimed in one of claims 34 or 35, characterised in that itincludes means (CFM2) for correcting the phase of each pulse of thereceived signal by a linearly increasing value the slope of which isrelated to a predefined frequency offset value. 38) Receiver as claimedin one of claims 34 or 37, characterised in that each decision valuecomputed at a time instant related to an output stream, respectively acombination of output streams, is the squared norm of said outputstream, respectively the combined output stream, evaluated at said timeinstant. 39) Receiver as claimed in one of claims 34 or 38,characterised in that each decision value computed at a time instantrelated to an output stream, respectively a combination of outputstreams, is a correlation merit factor defined by the ratio of theenergy at said time instant of said output stream, respectively saidcombination of output streams, divided by the energy of said outputstream, respectively said combination of output streams, averaged on twotime intervals defined respectively before and after said time instant.40) Receiver as claimed in one of claims 34 or 39, characterised in thatthe resulting stream combiner (RSC) is a bunch of correlators (COj)which are each intended to correlate at a given time instant a resultingstream with a comb of pulses related to a weighting code. 41) Receiveras claimed in one of claims 34 or 39, the set of I weighting codes is aset of orthogonal codes from which a fast transform is obtained,characterised in that the resulting stream combiner (RSC) includes meansfor processing L pulses of a resulting stream separated to each other byN positions with said fast transform. 42) Receiver as claimed in one ofclaims 34 or 39, characterised in that the resulting stream combiner(RSC) includes means for computing at a time instant the square root ofthe sum of the energy of L pulses separated by N positions from eachother of said resulting stream. 43) Receiver as claims in claim 42, eachof (L−1) components following the first component of each of said Iweighting codes being differentially encoded from its first component,the signal frame being possibly corrected in frequency, L peaksseparated by N positions from each other of a single resulting streamobtained from said maximal decision value and related to each of said atleast one antenna are considered, the first of said L peaks beinglocated at the time instant at which the signal frame is synchronised intime, characterised in that it further includes a weighting coderetriever (WCR) which includes means for obtaining a soft estimate of acomponent of rank q of the weighting code carried by the received signalframe from the product of the sum of the phase rotation and amplitudecorrected peaks of rank q of said at least one single resulting streamby a weighting value. 44) Receiver as claimed in one of claims 34 to 43,characterised in that it further includes means for correcting the phaseof each pulse of a received signal frame by a linearly increasing valuethe slope of which is obtained from a weighted average of at least Pslope estimations obtained for each of said at least one antenna, eachof said P slope estimations being weighted by an estimate of the squaredamplitude of one of said P path coefficients, the weighting value ofsaid average being the sum of the squared amplitude of said P pathcoefficients. 45) Signal frame transmitted from a transmitter of atelecommunication system to a receiver adapted to synchronise saidsignal frame from a synchronisation sequence (x^(i)(k)) included in saidsignal frame, characterised in that the synchronisation sequence(x^(i)(k)) is formed by the concatenation of a first and a second burstsof L complementary sequences, said first burst of L sequences beingobtained by the concatenation of L times a first N pulses complementarysequence (A) of the pair of complementary sequences ((A,B)), said secondburst of L sequences being obtained by the concatenation of L times asecond N pulses complementary sequence (B) of said pair of complementarysequences, the pulses of each complementary sequence (A) of rank q ofthe first burst of L sequences being multiplied by the component of rankq of a L components long weighting code (MC^(i)) belonging to a set of Iweighting codes known by the receiver beforehand, and the pulses of eachsequence of rank q of the second burst of L sequences being multipliedby the same component of rank q of said weighting code (MC^(i)). 46)Signal frame as claimed in claim 45, characterised in that saidsynchronisation sequence (x^(i)(k)) further includes at least one guardinterval, defined by a W bits long cyclic extension of either the firstor second complementary sequence, said at least one guard interval beinglocated either at the beginning or at the end of the first burst of Lsequences when said at least one guard interval is defined from thefirst complementary sequence (A), and said at least one guard intervalbeing located either at the beginning or at the end of the second burstof L sequences when said at least one guard interval is defined from thesecond complementary sequence (B). 47) Signal frame as claimed in one ofclaims 45 or 46, the weighting code (MC^(i)) belonging to a set of Iweighting codes, characterised in that the set of I weighting codes is aset of orthogonal codes from which a fast transform is obtained. 48)Signal frame as claimed in one of claims 45 to 47, characterised in thatthe weighting codes are multiplied by a same scrambling code. 49)Telecommunication system including at least one transmitter as claimedin one of claims 27 to 33, said at least one transmitter being intendedto transmit a signal frame to at least one receiver of saidTelecommunication system. 50) Telecommunication system as claimed inclaim 49, characterised in that said at least one receiver is defined asclaimed in one of claims 34 to 44.